Frequency shaping and adaptive rounding for O-QPSK and MSK transmission in polar coordinates

ABSTRACT

Systems and methods are directed to phase modulation of polar coordinates in a transmitter of wireless signals, to achieve high transmit power levels while meeting spectral mask and EVM requirements. An input signal is mapped to a sequence of modulation frequency (e.g., O-QPSK to MSK) to generate a mapped signal. A digital frequency shaping filter is applied to the mapped signal to generate a shaped signal. An adaptive rounding algorithm is applied to the shaped signal to generate a reduced bit-width signal. A digital frequency synthesizer is applied to the reduced bit-width signal to generate an analog waveform for transmission.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present Application for Patent claims the benefit of ProvisionalPatent Application No. 62/314,943 entitled “FREQUENCY SHAPING ANDADAPTIVE ROUNDING FOR O-QPSK AND MSK TRANSMISSION IN POLAR COORDINATES”filed Mar. 29, 2016, and assigned to the assignee hereof and herebyexpressly incorporated herein by reference in its entirety.

FIELD OF DISCLOSURE

Disclosed aspects relate to a transmitter of wireless signals. Morespecifically, exemplary aspects are directed to frequency shaping andadaptive rounding for transmission of offset-quadrature phase shiftkeying (O-QPSK) and minimum-shift keying (MSK) modulated signals inpolar coordinates.

BACKGROUND

Wireless communication systems may include transmitters and receivers(or combinations thereof) of wireless signals. In the case oftransmission of the wireless signals, a transmitter may modulate datasignals to be transmitted on to a carrier wave and a receiver mayreceive the modulated signals and demodulate the data signals. Varioustypes of modulation techniques are known in the art, such as phasemodulation, amplitude modulation, etc. In this disclosure, phasemodulation is considered in more detail.

Phase modulation refers to a type of modulation where data signals (orinformation) are digitally encoded as variations in an instantaneousphase of the carrier wave. In the context of digital signaltransmission, phase modulation is seen to switch between differentphases. Thus, phase modulation is generally referred to as phase shiftkeying (PSK). Numerous types of PSK are known in the art, such as,quadrature PSK (QPSK), offset-QPSK (O-QPSK), binary PSK (BPSK),minimum-shift keying (MSK), etc. It is possible to switch betweendifferent types of PSK.

For example, considering a time-domain implementation of a QPSKmodulator, an input bit stream of the data signals to be transmitted issplit into in-phase (I) and quadrature (Q) waveforms, which are thenseparately modulated by two carriers which are in phase quadrature(e.g., a sine and a cosine carrier wave which are varied in phase, whilekeeping amplitude and frequency constant). This allows transmission oftwo bits in each modulation symbol, with four possible different symbolssince the phase of the carrier wave can take on four possible values(e.g., 0, π/2, π, 3π/2), wherein each phase corresponds to a differentsymbol. It is seen that each modulated signal in QPSK can be representedas a BPSK signal and summed up to produce the QPSK signal. In anotherexample, while it is possible to generate an O-QPSK waveform in asimilar manner as described above for QPSK, by generating the I and Qwaveforms separately using I and Q modulators, in the time-domain,O-QPSK modulation can also be achieved by generating time-domainbaseband I and Q waveforms according to QPSK signaling followed by usinghalf-sine (HS) shaping filters, and shifting the Q waveform by half asymbol period with respect to the I waveform. As yet another example, anMSK modulator can be implemented by recognizing that the differencebetween O-QPSK and MSK lies in the way the input bits are mapped.

Accordingly it is seen that for various types of PSK signaling intime-domain, a transmitter can be implemented using I and Q modulators.Doing so makes it possible for the modulated signals to satisfy a“spectral mask”, which defines a power spectrum according to wirelesscommunication standards or regulations. For example, the spectral maskmay be satisfied by implementing a pair of digital low-pass filtersdesigned to suppress side-lobes of the I and Q modulated signals intheir signal power spectrum. In time-domain, finding filter coefficientsthat achieve both satisfactory side-lobe suppression and satisfactoryerror vector magnitude (EVM) is relatively straightforward, andtherefore, transmitters which implement time-domain PSK signaling can bedesigned to meet the spectral mask and EVM using conventional approachesknown in the art.

With the exploration of low-cost RF communication (e.g., WiFi,Bluetooth, Bluetooth Low Energy (BLE), etc.) seen in recent times, forexample, in emerging markets such as Internet-of-Things (IoT),frequency-domain PSK signaling is recognized as a better alternative totime-domain PSK signaling, since implementations of transmitters usingfrequency-domain signaling can incur less costs in comparison totransmitters using time-domain signaling.

For example, O-QPSK and MSK signals can also be generated using afrequency synthesizer, rather than the separate I and Q modulators asdiscussed above in the time-domain. In the frequency-domain, O-QPSK andMSK modulation can involve mapping the data signals or information bitsto corresponding waveforms in the frequency domain and feeding themapped symbols to a frequency synthesizer which generates a frequencymodulated (FM) signal. A modulator using the frequency synthesizer canbe designed with less area and can consume less power in comparison tothe I and Q modulators. However, unlike the straightforward case of Iand Q modulators in the time-domain, the spectral mask and EVMrequirements in the frequency domain are more difficult to satisfy, asthese metrics are related to standards specified for the low-costapplications discussed above (e.g., the well-known IEEE 802.15.4standard). At high transmit power levels (which can be desirable in manyscenarios in this application space), conventional frequency-domainimplementations of O-QPSK and MSK modulation, for example, with analogfrequency synthesizers, are seen to violate or exceed the specifiedspectral mask in an attempt to meet the EVM requirements.

Thus, there is recognized a need in the art for designs of transmittersin the frequency domain which can meet spectral mask and EVMrequirements for various modulation schemes such as O-QPSK or MSK in thefrequency domain, while retaining the desirable characteristics of lowcost and low power.

SUMMARY

The following presents a simplified summary relating to one or moreaspects disclosed herein. Example systems and methods are directed tophase modulation of polar coordinates in a transmitter of wirelesssignals, to achieve high transmit power levels while meeting spectralmask and EVM requirements. An input signal is mapped to a sequence ofmodulation frequency (e.g., O-QPSK to MSK) to generate a mapped signal.A digital frequency shaping filter is applied to the mapped signal togenerate a shaped signal. An adaptive rounding algorithm is applied tothe shaped signal to generate a reduced bit-width signal. A digitalfrequency synthesizer is applied to the reduced bit-width signal togenerate an analog waveform for transmission.

For example, an exemplary aspect is directed to a method for generatingphase modulated signals in polar coordinates for transmission in atransmitter of wireless signals, the method comprising mapping an inputsignal to a sequence at modulation frequency to generate a mappedsignal, applying a digital frequency shaping filter to the mapped signalto generate a shaped signal, applying an adaptive rounding algorithm tothe shaped signal to generate a reduced bit-width signal; and applying adigital frequency synthesizer to the reduced bit-width signal togenerate an analog waveform for transmission.

Another exemplary aspect is directed to a transmitter of wirelesssignals, comprising a mapping block configured to map an input signal inpolar coordinates to a sequence at modulation frequency to generate amapped signal, a digital frequency shaping filter configured to shapethe frequency of the mapped signal to generate a shaped signal, anadaptive rounding block configured to perform adaptive rounding of theshaped signal to generate a reduced bit-width signal, and a digitalfrequency synthesizer configured to generate an analog waveform fortransmission from the reduced bit-width signal.

Another exemplary aspect is directed to an apparatus configured fortransmission of wireless signals, the apparatus comprising means formapping an input signal in polar coordinates to a sequence at modulationfrequency to generate a mapped signal, means for digital shaping thefrequency of the mapped signal to generate a shaped signal, means foradaptively rounding the shaped signal to generate a reduced bit-widthsignal, and means for generating an analog waveform for transmissionfrom the reduced bit-width signal.

Yet another exemplary aspect is directed to a non-transitory computerreadable storage medium comprising code, which, when executed by aprocessor, causes the processor to perform operations for generatingphase modulated signals in polar coordinates for transmission ofwireless signals, the non-transitory computer readable storage mediumcomprising code for mapping an input signal to a sequence at modulationfrequency to generate a mapped signal, code for applying a digitalfrequency shaping filter to the mapped signal to generate a shapedsignal, code for applying an adaptive rounding algorithm to the shapedsignal to generate a reduced bit-width signal, and code for applying adigital frequency synthesizer to the reduced bit-width signal togenerate an analog waveform for transmission.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings are presented to aid in the description ofaspects of the invention and are provided solely for illustration of theaspects and not limitation thereof.

FIG. 1 illustrates a conventional transmitter system for transmission ofpolar coordinates.

FIG. 2 an exemplary transmitter system for transmission of polarcoordinates.

FIG. 3 illustrates waveforms for various phase modulation techniques.

FIGS. 4A-E illustrate aspects of digital frequency shaping filters.

FIG. 5A illustrates input-output plots to show accumulated quantization.

FIG. 5B illustrates a flow-chart of an adaptive rounding algorithm forreducing quantization error according to aspects of this disclosure.

FIG. 6 illustrates an example wireless transceiver with transmitter sideprocessing, according to aspects of the disclosure.

FIG. 7 illustrates example wireless devices, according to aspects of thedisclosure.

FIG. 8 illustrates an example method for phase modulation of polarcoordinates in a transmitter of wireless signals, according to aspectsof the disclosure.

DETAILED DESCRIPTION

Various aspects are disclosed in the following description and relateddrawings directed to specific aspects of the invention. Alternateaspects may be devised without departing from the scope of theinvention. Additionally, well-known elements of the invention will notbe described in detail or will be omitted so as not to obscure therelevant details of the invention.

The word “exemplary” is used herein to mean “serving as an example,instance, or illustration.” Any aspect described herein as “exemplary”is not necessarily to be construed as preferred or advantageous overother aspects. Likewise, the term “aspects of the invention” does notrequire that all aspects of the invention include the discussed feature,advantage or mode of operation.

The terminology used herein is for the purpose of describing particularaspects only and is not intended to be limiting of aspects of theinvention. As used herein, the singular forms “a”, “an”, and “the” areintended to include the plural forms as well, unless the context clearlyindicates otherwise. It will be further understood that the terms“comprises”, “comprising”, “includes”, and/or “including”, when usedherein, specify the presence of stated features, integers, steps,operations, elements, and/or components, but do not preclude thepresence or addition of one or more other features, integers, steps,operations, elements, components, and/or groups thereof.

Further, many aspects are described in terms of sequences of actions tobe performed by, for example, elements of a computing device. It will berecognized that various actions described herein can be performed byspecific circuits (e.g., application specific integrated circuits(ASICs)), by program instructions being executed by one or moreprocessors, or by a combination of both. Additionally, these sequence ofactions described herein can be considered to be embodied entirelywithin any form of non-transitory computer-readable storage mediumhaving stored therein a corresponding set of computer instructions thatupon execution would cause an associated processor to perform thefunctionality described herein. Thus, the various aspects of theinvention may be embodied in a number of different forms, all of whichhave been contemplated to be within the scope of the claimed subjectmatter.

Exemplary aspects of this disclosure are directed to transmitters ofwireless signals, configured to satisfy spectral mask and EVMrequirements for phase modulated signals. For example, exemplarytransmitters are designed to generate phase modulated symbols such asO-QPSK or MSK signals, (e.g., as specified in standards such as IEEE802.15.4 PHY) in frequency-domain (or “polar coordinates”) and for hightransmit power levels with analog frequency synthesizers. Accordingly,aspects of an exemplary transmitter design comprise a discrete-time ordigital frequency shaping filter and an adaptive rounding algorithm, asfurther explained below.

The frequency shaping filter is configured to lower or suppress thesignal side-lobes in frequency while maintaining the EVM at a low level,wherein the frequency shaping filter can be implemented as a polyphasefilter with low hardware costs in some examples.

The adaptive rounding algorithm, which can be implemented in an adaptiverounding block, for example, is configured to prevent the inputfrequency to the analog frequency synthesizer from having bias, sincesuch a bias would only have been introduced after quantizing thefrequency shaping filter output to the frequency synthesizer input bitwidth. The adaptive rounding algorithm is also configured to prevent thephase of the modulated signal from drifting in an unbounded manner,wherein it is noted that if the phase drift is left unchecked, it canlead to large, unbounded growth in EVM.

Accordingly, the combination of the digital frequency shaping filter andthe adaptive rounding algorithm can ensure that the spectral mask issatisfied while meeting the EVM requirements.

It will be understood that although O-QPSK or MSK schemes may bediscussed herein as representative phase modulation schemes in exemplaryaspects, the disclosure is not limited to these schemes. As such thescope of exemplary aspects can be easily extended to any other suitablemodulation scheme based on this disclosure.

With reference to FIG. 1, a conventional implementation of phasemodulation in polar coordinates will first be discussed. Illustrated inFIG. 1 are aspects of a phase modulator in transmitter 100 of wirelesssignals (e.g., a transmitter of RF signals according to IEEE 802.15.4O-QPSK PHY). Block 102 receives a data stream of input signals 101 andapplies phase modulation techniques such as O-QPSK or MSK in thefrequency-domain or polar coordinates, to generate output 103 comprisingmapped symbols according to the PSK scheme applied in block 102. Output103 is supplied to frequency synthesizer 104 to generate frequencymodulated (FM) signal 105. As previously discussed, FM signal 105 mayviolate the spectral mask for the applicable standards governingtransmission of wireless signals by transmitter 100. It is recognizedthat FM signal 105 is an analog signal, and so analog filter 106 isapplied to FM signal 105, e.g., for side-lobe suppression in FM signal105, in an effort to meet the spectral mask. However, analog filter 106is expensive, since it implements analog components, andcorrespondingly, also consumes significant area and power. Furthermore,the implementation of analog filter 106 is not easily scalable and somay be unsuitable for high transmit power of frequency shaped signal 107generated at the output of analog filter 106.

With reference now to FIG. 2, exemplary transmitter 200 is illustrated,wherein transmitter 200 is configured with exemplary phase modulationtechniques. Transmitter 200, as will be further explained below, can beconfigured to ensure that EVM requirements are met while also meetingspectral mask specifications even at high transmit power levels whileapplying phase modulation in polar coordinates. In exemplary aspects,transmitter 200 avoids the need for an analog filter such as analogfilter 106 of FIG. 1. In exemplary aspects, transmitter 200 implements adigital frequency shaping filter and an adaptive rounding algorithm asdiscussed below.

Referring to FIG. 2, input signal 201 is received by transmitter 200.Input signal 201 may comprise a data stream. Since multiple symbols maybe transmitted per data bit, based on the type of PSK applied, themultiple symbols per data bit are referred to as “chips”, such that thesymbol rate of input signal 201 is referred to as a chip rate. As shown,input signal 201 may have a chip rate of 1 bit, but this is merelyexemplary and not a limitation. In block 202, a chosen PSK is applied tomap input signal 201 to the phase-modulated output 203. Although block202 can be designed for a specific type of PSK, it is possible tointerchange or map between the different PSK algorithms. This isexplained for the case of O-QPSK to MSK mapping with a brief departurefrom FIG. 2 to FIG. 3.

Referring to FIG. 3, table 300 is shown to illustrate that QPSK/O-QPSKchips can be mapped to MSK chips. Specifically, waveforms 306, 308, 310,and 312 are shown in the time-domain. For example, waveform 310 showsphase relations for I and Q streams for an example input signal intime-domain. Waveform 306 shows QPSK modulated I and Q streamscorresponding to waveform 310. Waveform 308 shows how the O-QPSKwaveform is obtained by shifting the Q waveform of waveform 306 by halfa symbol period with respect to the I waveform. Waveform 312 illustratesan MSK phase trellis corresponding to waveform 310.

With continuing reference to FIG. 3, rows 302, 304, and 314 show examplemappings in the frequency-domain. For example, row 302 provides a serialnumber and row 304 shows QPSK chips c(k) (which map to waveform 306)corresponding to the serial numbers in row 302. MSK chips d(k) are shownin row 314, wherein it is observed that the serial numbers of row 302corresponding to MSK chips in 314 can be easily obtained by shifting theQPSK chips of row 304 by one symbol period, which, incidentally alsocorresponds to O-QPSK chips (not shown) based on the above explanation.

Thus, returning to FIG. 2, it is seen that if block 202 is designed forMSK, then QPSK or O-QPSK chips can be easily mapped to the MSK chips.Accordingly, an illustrative description of exemplary features will beprovided with MSK mapping applied in block 202, keeping in mind that anyother type of PSK can be equally applicable to the exemplary aspects,based on straightforward mapping between the different types of phasemodulation. Accordingly, with MSK mapping considered in an example,without loss of generality, block 202 performs MSK mapping on inputsignal 201 to generate mapped signal 203. Mapped signal 203 is also adigital signal, which has the same chip rate of input signal 201 (i.e.,a chip rate of 1 bit in this example). Rather than feed mapped signal203 to a frequency synthesizer (as in the conventional case discussedwith reference to FIG. 1), in exemplary transmitter 200, mapped signal203 is provided to a digital frequency shaping filter 204.

Frequency shaping filter 204 is designed to help suppress side-lobes. Ingeneral, suppression of side-lobes may lead to an increase in EVM.Therefore, frequency shaping filter 204 is designed to ensure that EVMdoes not significantly increase. Furthermore, frequency shaping filter204 causes a quantized input to be provided to frequency synthesizer208. Quantization of the input to frequency synthesizer 208 may alsolead to an increase in EVM (referred to herein, as the “quantizationeffect”). To avoid an increase in EVM due to the quantization effect,the aforementioned adaptive rounding algorithm is implemented inadaptive rounding block 206, which will be discussed in further detailin the following sections.

Example implementations of frequency shaping filter 204 will bediscussed with relation to FIGS. 4A-E in the following sections. Thefunctionality of frequency shaping filter 204 will now be discussed inmore detail.

Considering, for the sake of explanation, specifications of an examplestandard such as the IEEE 802.15.4 PHY (hereafter “the standard”) areconsidered. When operating in the industrial, scientific and medical(ISM) 2450 MHz radio band, the standard defines that signal transmissionshould conform to the table identified as Table 1 below:

TABLE 1 IEEE spectral mask requirements Frequency Relative limitAbsolute limit |f − f_(c)| > 3.5 MHz −20 dB −30 dBmFrom Table 1 (and keeping in mind that for both relative and absolutelimits, the average spectral power is to be measured using a 100 KHzresolution bandwidth) for the relative limit, the reference level isspecified to be the highest average spectral power measured within ±1MHz of the carrier frequency.

Accordingly, for O-QPSK, the O-QPSK power spectral density is providedby

${{\Phi_{XX}(f)} = {\frac{G}{2\pi^{2}}{\left( \frac{\cos\left( {\pi\;{fT}_{s}} \right)}{\left( {fT}_{s} \right)^{2} - \frac{1}{4}} \right)^{2}\left\lbrack {W\text{/}{Hz}} \right\rbrack}}},$where T_(s)=1 μs.

As explained previously, it is desirable for transmitter 200 to transmitsignals at high transmit power levels, e.g., above the relative limit of20 dBm. Thus, considering a transmit power of 21 dBm, as an example of atransmit power to achieve, and setting G=1.2978×10⁻⁷, with 100 KHzresolution bandwidth (RBW), the following equations are reached fortransmit power levels “P” below:P=+21.1 dBm(100%), |f−f _(c)|<20 MHzP=+21.0 dBm(97.0%), |f−f _(c)|<1.0 MHzP=−12.9 dBm(0.04%), |f−f _(c)|>3.5 MHz

As seen, these transmit power levels do not meet the requirements forthe absolute power limit shown in Table 1, i.e., −30 dBm, forfrequencies beyond 3.5 MHz, as per the standard. In addition, theFederal Communications Commission (FCC) requirement for the absolutepower limit for the top of the ISM band is −41 dBm/MHz for frequencyranges between 2483.5 MHz to 2500 MHz. Thus, even if the last channel,(i.e., channel 26 at the 2480 MHz band) is not used for transmission,the previous channel (i.e., the channel at 2475 MHz band) should be ableto meet the absolute power limit requirement. This means that at least17 dB of side-lobe suppression is needed to meet the spectral maskdefined in the standard, and even more suppression to meet the FCCrequirement. Furthermore, spectral regrowth due to nonlinear phaseamplification (PA) also needs to be accounted for. To satisfy thesevarious requirements, frequency shaping filter 204 is designed toimplement shaping via a frequency deviation (fdev) waveform for MSK.

To meet the mask requirement the MSK fdev waveform, which is the timederivative of the MSK phase trellis shown in row 312 of FIG. 3, isassumed to be an ideal pulse train. Frequency shaping filter 204 canthus be configured to shape the fdev waveform, by implementing, forexample, a filter which can reduce the power spectral density (PSD)side-lobe level. Incidentally, this would also enable reduction by about35% in the EVM, from the EVM defined as acceptable in the standard.

As previously mentioned, the error-vector magnitude (EVM) determines themodulation accuracy of the transmitter. To calculate the EVM a timerecord of N received complex chip values (Ī_(j), Q _(j)) is captured.The error vector is the distance from the ideal position to the actualposition, represented by the equation (Ī_(j), Q _(j))=(I_(j),Q_(j))+(δI_(j), δQ_(j)). The EVM is correspondingly defined as

${EVM}\overset{\Delta}{=}\sqrt{\frac{\frac{1}{N}{\sum\limits_{j = 1}^{N}\left( {{\delta\; I_{j}^{2}} + {\delta\; Q_{j}^{2}}} \right)}}{\frac{1}{N}{\sum\limits_{j = 1}^{N}\left( {I_{j}^{2} + Q_{j}^{2}} \right)}}}$

According to the standard, the O-QPSK PHY is required to have EVM valuesless than 35% when measured over 1000 chips. In exemplary aspects, it isrecognized that frequency shaping filter 204 can be implemented using afamily of truncated raised-cosine filters in order to satisfy the abovespectral mask requirements and also display very low EVMcharacteristics. Referring to FIG. 4A, a square raised cosine filter 402with a roll off factor of 0 (i.e., a sinc filter) is shown for oneimplementation of frequency shaping filter 204. In FIG. 4A, it is seenthat although the spectral side-lobes get suppressed, at the same timeEVM goes up as the impulse response main lobe is stretched to span atime duration longer than 1 chip period. Thus, the effect of the risingEVM needs to be combated.

For an efficient implementation of filter 402 in FIG. 4A, an N-tapeven-symmetric filter with z-domain transfer function is provided asfollowsH(z)=h(0)+h(1)z ⁻¹ + . . . +h(N−1)z ^(−(N-1))

For a direct-form implementation of filter 402, the characteristic ofH(z) having an even symmetry is exploited, wherein, by re-arranging thecoefficients of H(z) in the above equation, the following representationof H(z) is obtained:H(z)=h(0)(1+z ^(−(N-1)))+h(1)(z ⁻¹ +z ^(−(N-2)))+ . . . +h(N/2−1)(z^(−N/2-1) z ^(−N/2))

Considering the case of filter 402 implemented as a 3-chip-longtruncated sinc shaping filter, and assuming that the transmission chiprate is 2 MHz and transmission sample rate is 26 MHz, the 2 MHz chipsneed to be repeated 13 times and the filter will have 3×13=39 taps. Tomake the filter even-symmetric, one more tap is added, to obtain N=40taps. FIG. 4B is a plot 404 of all the filter taps. It is noted thatonly 20 of the tap values need to be stored. Also, since samples of theinput signal 201 are either +1 or −1 in FIG. 2, there is no need foractual multiplication. A structural implementation 406 of filter 402 isshown in FIG. 4C.

Another implementation of frequency shaping filter 204 can be apolyphase structure. For a polyphase structure, a sample repetitionwithin a chip can be performed as explained above, keeping in mind thatsample repetition within a chip is equivalent to up-sampling (zeropadding) followed by convolution with a square pulse that has a durationof the chip. Thus, when the square pulse is convoluted with thetruncated sinc filter, as shown in plot 408 of FIG. 4D, because of theconvolution, plot 408 is one chip duration longer, i.e., there are 13more taps (total of 52 taps).

Referring to plot 408 of FIG. 4D, the frequency shaping to beimplemented in frequency shaping filter 204 may be recast intoup-sampling of mapped signal 203 followed by interpolation. Therefore, apolyphase filter 410 as shown in FIG. 4E, using interpolating filterswhere the entire filter can run at the input chip rate, i.e., 2 MHz,instead of the output sample rate, i.e., 26 MHz may be used. In thisregard, an N tap even-symmetric filter with the below transfer functionis used as a starting point:

${H(z)} = {\frac{Y(Z)}{X(Z)} = {{h(0)} + {{h(1)}z^{- 1}} + \ldots\; + {{h\left( {N - 1} \right)}z^{{- N} + 1}}}}$Wherein, it is assumed that N=LM where M is the interpolation factor,i.e.,

$M = {\frac{26\mspace{14mu}{MHz}}{2\mspace{14mu}{MHz}} - 13.}$Since N=52 is the total number of taps in the interpolation filter, Lbecomes 4.

The above transfer function can be recast in the following manner:

  H(z) = H₀(z^(M)) + z⁻¹H₁(z^(M)) + z^(−(M − 1))H_(M − 1)(z^(M)) + …   where,  H₀(z^(M)) = h(0) + h(M)z^(−M) + …  + h((L − 1)M)z^(−(L − 1)M)  H₁(z^(M)) = h(1) + h(M + 1)z^(−M) + … + h((L − 1)M + 1)z^(−(L − 1)M)  ⋮            ⋮H_(M − 1)(z^(M)) = h(M − 1) + h(2M − 1)z^(−M) + … + h(LM − 1)z^(−(L − 1)M),

FIG. 4E shows the recast transfer function implemented using thestructure of polyphase filter 410 a. Polyphase filter 410 a isequivalently modified to the structure of polyphase filter 410 b, andusing Noble identity, to polyphase filter 410 c, to eventually reach thestructure of polyphase filter 410 d. Although any polyphase filterimplementation can be used for frequency shaping filter 204 of FIG. 2,polyphase filter 410 d may be used in exemplary aspects. Polyphasefilter 410 d is seen to effectively reduce power consumption (side-lobesuppression) to meet the PSD, and also keeps EVM from increasing, whichallows EVM requirements to be met.

Returning now to FIG. 2, regardless of the specific implementationchosen for frequency shaping filter 204, frequency shaped output 205 offrequency shaping filter 204 may be quantized, e.g., represented as anN-bit wide signal based on the sample rate applied. The quantizedfrequency shaped output 205 is fed to adaptive rounding block 206.Adaptive rounding block 206 is configured to apply an adaptive roundingalgorithm designed to prevent phase drift and curb EVM growth due to theaforementioned quantization effect. The adaptive rounding algorithm andexemplary implementations of adaptive rounding block 206 will now beexplained.

An aspect of adaptive rounding block 206 is directed to reducing the bitwidth of the digital signal 207 (e.g., from N-bits to N-M bits as shownin FIG. 2) which is provided to frequency synthesizer 208, as this wouldreduce the area, power consumption and design complexity of frequencysynthesizer 208. However, reducing the bit width can lead to a biggerquantization error and EVM growth. Even if large bit widths are allowedfor internal signals of transmitter 200, the output of transmitter 200,which is an instantaneous modulation frequency, may need to be quantizedbefore it is fed to the frequency synthesizer. However, it is alsoobserved that in the case of a transmitter designed according to theaforementioned IEEE 802.15.4 standard, that a majority of the filtercoefficient sets for frequency shaping filter 204 are seen to display askewed quantization error distribution. In other words, quantizationerrors tend to be mostly positive or mostly negative.

Referring to FIG. 5A, output vs input waveforms 502, 504, and 506showing quantization error are illustrated. Waveform 502 refers to anormal condition where there is no accumulated quantization error andthe reduced bit width output 207 (e.g., N-M bits) of adaptive roundingblock 206 follows its input 207 (e.g., N-bits) in a staircase function.However, if there is a positive quantization error, waveform 504illustrates the output drifting away in the positive direction from theinput, while if there is a negative quantization error, waveform 506illustrates the output drifting away in the negative direction from theinput. This means that an accumulated quantization error (waveforms504/506) for the instantaneous modulation frequency would cause thephase of output 209 of frequency synthesizer 208 to drift in anunchecked manner, which can in turn cause the EVM to deteriorate.

To prevent the phase of the output from drifting, the adaptive roundingalgorithm is designed to monitor the accumulated quantization error andadapt a quantization threshold such that future quantization erroroccurs in the desired direction (the opposite direction of theaccumulated quantization error at a given instant) more easily.

With reference to FIG. 5B, a flow-chart 550 pertaining to an exampleadaptive rounding algorithm according to this disclosure will beexplained. In FIG. 5B, floor(.) is an operator which returns the largestinteger no greater than the argument, whereinoutput=floor(input+bias+0.5).

In block 552, accumulated quantization error is initialized asaccumulated_quantization_error=accumulated_quantization_error+input−outputIn decision block 554,if (accumulated_quantization_error>threshold_big_positive)then in block 556 the bias is changed in the positive direction:bias=bias_positiveOtherwise, the flow proceeds to decision block 558, whereinif (accumulated_quantization_error<threshold_big_negative)then in block 560 the bias is changed in the negative direction:bias=bias_negativeOtherwise, the flow proceeds to decision block 562, whereinif (accumulated_quantization_error<threshold_normal_positive ANDaccumulated_quantization_error>threshold_normal_negative)then in block 564 the bias is set to zero:bias=0Otherwise, flow-chart 550 ends in block 566, where there is no change inthe bias.

As seen from flow-chart 550 of FIG. 5B, by adaptively changing the biasdepending on how the accumulated quantization error compares to a set ofthresholds: (e.g., big positive threshold, big negative threshold,normal positive threshold, and normal negative threshold in the abovepseudo-code), the phase error is prevented from drifting. The aboveflow-chart 550 or equivalent pseudo-code may be implemented by anysuitable hardware or combination of hardware/software in adaptiverounding block 206. For example, the adaptive rounding block can includea rounding block configured to generate a quantized value based on theinput sample value and a bias, a logic block configured to compute theaccumulated quantization error, and a logic block configured todetermine a rounding bias based on the current computed quantizationerror and a previous accumulated quantization error. As such, a hardwareimplementation of adaptive rounding block 206 would not incursignificant area, or power but can enables a significant area and powersavings in transmitter 200 by reducing the bit width of the input 207 tofrequency synthesizer 208 (i.e., the output of adaptive rounding block206 which can be reduced to a sample rate of N-M bits). In this manner,frequency synthesizer 208 can be designed with smaller number of inputbits while meeting EVM requirements.

It is observed that if more simplistic approaches were relied on ratherthan the exemplary adaptive rounding algorithm, such as adding orsubtracting a fixed value when the accumulated quantization errorbecomes significant in an effort to reduce the drift, such abruptcorrection techniques may cause side-lobe power spectral density torise, which is undesirable. In contrast, the adaptive rounding algorithmis designed to apply a more smooth reduction in the accumulatedquantization error, thus allowing for better side-lobe suppression whilemeeting EVM requirements.

FIG. 6 illustrates an example wireless transceiver 600 according toaspects of the disclosure. The illustrated example of wirelesstransceiver 600 includes PLL 602, modulator 604, digital controller 610,buffers 612 and 614, transmit amplifiers 616, transmit matching network618, transmit/receive switch 620, antenna 622, divider 624, receivematching network 626, front end amplifier 628, mixer 630, low passfilter 632, mixers 634 and 636, low pass filters 638 and 640, andanalog-to-digital converters (ADCs) 642 and 644. Wireless transceiver600 is illustrated as having distinct transmit and receive processingpaths. Exemplary aspects of this disclosure may be applicable to thetransmit processing path, as discussed in the above sections.

With reference now to FIG. 7, example wireless devices 700A and 700B,according to aspects of the disclosure are illustrated. In someexamples, wireless devices 700A and 700B may herein be referred to aswireless mobile stations. The example wireless device 700A isillustrated in FIG. 7 as a calling telephone and wireless device 700B isillustrated as a touchscreen device (e.g., a smart phone, a tabletcomputer, etc.). As shown in FIG. 7, an exterior housing 735A ofwireless device 700A is configured with antenna 705A, display 710A, atleast one button 715A (e.g., a PTT button, a power button, a volumecontrol button, etc.) and keypad 720A among other components, not shownin FIG. 7 for clarity. An exterior housing 735B of wireless device 700Bis configured with touchscreen display 705B, peripheral buttons 710B,715B, 720B and 725B (e.g., a power control button, a volume or vibratecontrol button, an airplane mode toggle button, etc.), at least onefront-panel button 730B (e.g., a Home button, etc.), among othercomponents, not shown in FIG. 7 for clarity. For example, while notshown explicitly as part of wireless device 700B, wireless device 700Bmay include one or more external antennas and/or one or more integratedantennas that are built into the exterior housing 735B of wirelessdevice 700B, including but not limited to WiFi antennas, cellularantennas, satellite position system (SPS) antennas (e.g., globalpositioning system (GPS) antennas), and so on.

While internal components of wireless devices such as the wirelessdevices 700A and 700B can be embodied with different hardwareconfigurations, a basic high-level configuration for internal hardwarecomponents is shown as platform 702 in FIG. 7. Platform 702 can receiveand execute software applications, data and/or commands transmitted froma radio access network (RAN) that may ultimately come from a corenetwork, the Internet and/or other remote servers and networks (e.g., anapplication server, web URLs, etc.). Platform 702 can also independentlyexecute locally stored applications without RAN interaction. Platform702 can include a transceiver 706 operably coupled to an applicationspecific integrated circuit (ASIC) 708, or other processor,microprocessor, logic circuit, or other data processing device. ASIC 708or other processor executes an application programming interface (API)710 layer that interfaces with any resident programs in a memory 712 ofthe electronic device. Memory 712 can be comprised of read-only orrandom-access memory (RAM and ROM), EEPROM, flash cards, or any memorycommon to computer platforms. Platform 702 also can include a localdatabase 714 that can store applications not actively used in memory712, as well as other data. Local database 714 is typically a flashmemory cell, but can be any secondary storage device as known in theart, such as magnetic media, EEPROM, optical media, tape, soft or harddisk, or the like.

In one aspect, wireless communications by wireless devices 700A and 700Bmay be enabled by the transceiver 706 based on different technologies,such as CDMA, W-CDMA, time division multiple access (TDMA), frequencydivision multiple access (FDMA), Orthogonal Frequency DivisionMultiplexing (OFDM), GSM, 2G, 3G, 4G, LTE, or other protocols that maybe used in a wireless communications network or a data communicationsnetwork. Voice transmission and/or data can be transmitted to theelectronic devices from a RAN using a variety of networks andconfigurations. Accordingly, the illustrations provided herein are notintended to limit the aspects of the invention and are merely to aid inthe description of aspects of aspects of the invention.

Accordingly, aspects of the present disclosure can include a wirelessdevice (e.g., wireless devices 700A, 700B, etc.) configured, andincluding the ability to perform the functions as described herein. Forexample, transceiver 706 may be implemented as wireless transceiver 600of FIG. 6, including the transmit processing path. As will beappreciated by those skilled in the art, the various logic elements canbe embodied in discrete elements, software modules executed on aprocessor or any combination of software and hardware to achieve thefunctionality disclosed herein. For example, ASIC 708, memory 712, API710 and local database 714 may all be used cooperatively to load, storeand execute the various functions disclosed herein and thus the logic toperform these functions may be distributed over various elements.Alternatively, the functionality could be incorporated into one discretecomponent. Therefore, the features of the wireless devices 700A and 700Bin FIG. 7 are to be considered merely illustrative and the invention isnot limited to the illustrated features or arrangement.

It will be appreciated that aspects include various methods forperforming the processes, functions and/or algorithms disclosed herein.For example, FIG. 8 illustrates an example method 800 for generatingphase modulated signals in polar coordinates, in a transmitter (e.g.,200) of wireless signals.

In block 802, method 800 comprises mapping an input signal (e.g., 201)to a sequence of modulation frequency (e.g., in block 202) to generate amapped signal (e.g., 203).

In block 804, method 800 comprises applying a digital frequency shapingfilter (e.g., frequency shaping filter 204) to the mapped signal togenerate a shaped signal (e.g., 205 of N bits).

In block 806, method 800 comprises applying an adaptive roundingalgorithm (e.g., in adaptive rounding block 206) to the shaped signal togenerate a reduced bit-width signal (e.g., 207), e.g., which satisfiesEVM requirements.

In block 806, method 800 comprises applying a digital frequencysynthesizer (e.g., frequency synthesizer 208) to the reduced bit-widthsignal to generate an analog waveform (e.g., 209, O-QPSK waveform) fortransmission, wherein the analog waveform, e.g., which satisfies aspecified spectral mask,

Further, those of skill in the art will appreciate that the variousillustrative logical blocks, modules, circuits, and algorithm stepsdescribed in connection with the aspects disclosed herein may beimplemented as electronic hardware or a combination of computer softwareand electronic hardware. To clearly illustrate this interchangeabilityof hardware and hardware-software combinations, various illustrativecomponents, blocks, modules, circuits, and steps have been describedabove generally in terms of their functionality. Whether suchfunctionality is implemented as hardware or software depends upon theparticular application and design constraints imposed on the overallsystem. Skilled artisans may implement the described functionality invarying ways for each particular application, but such implementationdecisions should not be interpreted as causing a departure from thescope of the present invention.

The methods, sequences and/or algorithms described in connection withthe aspects disclosed herein may be embodied directly in hardware, in asoftware module executed by a processor, or in a combination of the two.A software module may reside in RAM memory, flash memory, ROM memory,EPROM memory, EEPROM memory, registers, hard disk, a removable disk, aCD-ROM, or any other form of storage medium known in the art. Anexemplary storage medium is coupled to the processor such that theprocessor can read information from, and write information to, thestorage medium. In the alternative, the storage medium may be integralto the processor.

Accordingly, an aspect of the invention can include a non-transitorycomputer-readable media embodying a method for phase modulation of polarcoordinates in a transmitter of wireless signals. Accordingly, theinvention is not limited to illustrated examples and any means forperforming the functionality described herein are included in aspects ofthe invention.

While the foregoing disclosure shows illustrative aspects of theinvention, it should be noted that various changes and modificationscould be made herein without departing from the scope of the inventionas defined by the appended claims. The functions, steps and/or actionsof the method claims in accordance with the aspects of the inventiondescribed herein need not be performed in any particular order.Furthermore, although elements of the invention may be described orclaimed in the singular, the plural is contemplated unless limitation tothe singular is explicitly stated.

What is claimed is:
 1. A method for generating phase modulated signalsin polar coordinates for transmission in a transmitter of wirelesssignals, the method comprising: mapping an input signal to a sequence atmodulation frequency to generate a mapped signal; applying a digitalfrequency shaping filter to the mapped signal to generate a shapedsignal; applying an adaptive rounding algorithm to the shaped signal togenerate a reduced bit-width signal, wherein applying the adaptiverounding algorithm comprises: generating a quantized value from theshaped signal and a bias; computing an accumulated quantization error;and determining a rounding bias based on the computed accumulatedquantization error and a previous accumulated quantization error; andapplying a digital frequency synthesizer to the reduced bit-width signalto generate an analog waveform for transmission.
 2. The method of claim1, wherein the mapping comprises minimum-shift keying (MSK) oroffset-quadrature phase shift keying (O-QPSK) mapping.
 3. The method ofclaim 1, wherein applying the digital frequency shaping filter comprisesreducing side-lobes in frequency of the analog waveform fortransmission.
 4. The method of claim 1, wherein the digital frequencyshaping filter is a polyphase filter.
 5. The method of claim 1, whereinthe analog waveform for transmission meets error vector magnitude (EVM)spectral mask requirements according to IEEE 802.15.4 standards at hightransmit power levels.
 6. A transmitter of wireless signals, comprising:a mapping block configured to map an input signal in polar coordinatesto a sequence at modulation frequency to generate a mapped signal; adigital frequency shaping filter configured to shape the frequency ofthe mapped signal to generate a shaped signal; an adaptive roundingblock configured to perform adaptive rounding of the shaped signal togenerate a reduced bit-width signal, wherein the adaptive rounding blockis configured to: generate a quantized value from the shaped signal anda bias; compute an accumulated quantization error; and determine arounding bias based on the computed accumulated quantization error and aprevious accumulated quantization error; and a digital frequencysynthesizer configured to generate an analog waveform for transmissionfrom the reduced bit-width signal.
 7. The transmitter of claim 6,wherein the mapping block is configured to map the input signal based onminimum-shift keying (MSK) or offset-quadrature phase shift keying(O-QPSK).
 8. The transmitter of claim 6, wherein the digital frequencyshaping filter is configured to reduce side-lobes in frequency of theanalog waveform for transmission.
 9. The transmitter of claim 6, whereinthe digital frequency shaping filter is a polyphase filter.
 10. Thetransmitter of claim 6, wherein the analog waveform for transmission isconfigured to meet error vector magnitude (EVM) spectral maskrequirements according to IEEE 802.15.4 standards at high transmit powerlevels.
 11. An apparatus configured for transmission of wirelesssignals, the apparatus comprising: means for mapping an input signal inpolar coordinates to a sequence at modulation frequency to generate amapped signal; means for digital shaping the frequency of the mappedsignal to generate a shaped signal; means for adaptively rounding theshaped signal to generate a reduced bit-width signal comprising meansfor generating a quantized value from the shaped signal and a bias;means for computing an accumulated quantization error; and means fordetermining a rounding bias based on the computed accumulatedquantization error and a previous accumulated quantization error; andmeans for generating an analog waveform for transmission from thereduced bit-width signal.
 12. The apparatus of claim 11, comprisingmeans for mapping the input signal based on minimum-shift keying (MSK)or offset-quadrature phase shift keying (O-QPSK).
 13. The apparatus ofclaim 11, comprising means for reducing side-lobes in analog waveformfor transmission.
 14. The apparatus of claim 11, wherein the analogwaveform for transmission meets error vector magnitude (EVM) spectralmask requirements according to IEEE 802.15.4 standards at high transmitpower levels.
 15. A non-transitory computer readable storage mediumcomprising code, which, when executed by a processor, causes theprocessor to perform operations for generating phase modulated signalsin polar coordinates for transmission of wireless signals, thenon-transitory computer readable storage medium comprising: code formapping an input signal to a sequence at modulation frequency togenerate a mapped signal; code for applying a digital frequency shapingfilter to the mapped signal to generate a shaped signal; code forapplying an adaptive rounding algorithm to the shaped signal to generatea reduced bit-width signal, comprising: code for generating a quantizedvalue from the shaped signal and a bias; code for computing anaccumulated quantization error; and code for determining a rounding biasbased on the computed accumulated quantization error and a previousaccumulated quantization error; and code for applying a digitalfrequency synthesizer to the reduced bit-width signal to generate ananalog waveform for transmission.
 16. The non-transitory computerreadable storage medium of claim 15, wherein the mapping comprisesminimum-shift keying (MSK) or offset-quadrature phase shift keying(O-QPSK) mapping.
 17. The non-transitory computer readable storagemedium of claim 15, comprising code for reducing side-lobes in frequencyof the analog waveform for transmission.
 18. The non-transitory computerreadable storage medium of claim 15, wherein the analog waveform fortransmission meets error vector magnitude (EVM) spectral maskrequirements according to IEEE 802.15.4 standards at high transmit powerlevels.